1. Field of the Invention
The present invention relates to a switching power supply apparatus and, in particular, to such apparatus which operates with a relatively high efficiency.
2. Description of the Prior Art
In the prior art, high frequency distortion of power signals obtained from commercial a.c. power supplies occurs in visual equipment, such as, color televisions, television projectors and monitor equipment; video equipment, such as, video tape recorders and video disc players; audio equipment; office automation equipment and so forth. As a result, the so-called power efficiency of signals obtained from such a.c. power supplies is reduced or minimized.
In an attempt to improve the power efficiency, power choke coil inductors may be inserted in the a.c. line, or rectangular-wave shape converter circuits or active filter circuits employing oscillating wave shape converters may be utilized. However, these techniques are relatively expensive. Further, power choke coil inductors are relatively heavy and may cause problems due to magnetic flux leakage. Furthermore, the use of active filter circuits may also present problems relating to high electro-magnetic interference (EMI), power loss and the need of a relatively large area on a board, and so forth.
A partial rectification smoothing circuit having a voltage resonance converter circuit was therefore developed which can reduce high frequency distortion in signals obtained from a commercial a.c. power supply or the like, and provide power signals with a relatively high efficiency. Such circuit also produces relatively low EMI levels, and has a relatively small size, low weight and low cost.
FIG. 1 illustrates a circuit 100 which is an example of a switching power supply apparatus having a voltage resonance converter circuit. As shown therein, such circuit 100 generally includes an input circuit 101, a full wave rectification circuit 5, a partial rectification smoothing circuit 102 and a switching power supply circuit 12.
The input circuit 101 is coupled to the full wave rectification circuit 5 and generally includes a commercial a.c. power supply 1, a line filter transformer 4 and a capacitor C5. More specifically, line 2 is connected between one terminal of the power supply 1 and one input terminal of the rectification circuit 5 by way of winding 4a of the line filter transformer 4. Similarly, line 3 is connected between the other terminal of the power supply 1 and another input terminal of the rectification circuit 5 by way of winding 4b of the line filter transformer 4. The capacitor C5 is connected across lines 2 and 3 between the line filter transformer 4 and the full wave rectification circuit 5. The capacitor C5 and the line filter transformer 4 form a low-pass filter. The input circuit 101 is adapted to produce alternating voltage signals and to supply such signals to the full wave rectification circuit 5.
The full wave rectification circuit 5 is adapted to rectify the alternating voltage signals received from the input circuit 101 and to produce therefrom direct current pulsating voltage signals. The full wave rectification circuit 5 is formed of four diodes Da, Db, Dc and Dd which are connected together in a bridge-like manner. An output terminal on the negative side of the full rectification circuit 5 is connected to ground. An output terminal on the positive side of the full wave rectification circuit 5 is connected by way of a line 6 to a capacitor Ci and from there through a capacitor C1 to ground.
The partial rectification smoothing circuit 102 is connected to the line 6 and is adapted to smooth the d.c. pulse voltage signals supplied from the rectification circuit 5. Such circuit 102 includes the smoothing capacitor Ci and a voltage resonance converter 7. The voltage resonance converter 7 includes a converter transformer CT having a primary winding Nc1, a secondary winding Nc2 and another winding Ncb; a switching transistor Q1; and a resonance capacitor Cr which is connected in parallel across the collector and emitter of the switching transistor Q1. As a result, when the switching transistor Q1 is off, a parallel resonance voltage Vcp1 is generated by the capacitor Cr and the inductance of the primary winding Nc1 of the converter transformer CT. On the other hand, when the switching transistor Q1 is on, the collector current Icp1 is determined by the inductance of the primary winding Nc1 of the converter transformer CT, the input voltage V1, and the on time of transistor Q1.
A switching signal is generated by a first resonance circuit formed of a fixed time constant capacitor CB, the winding Ncb, and a fixed time constant inductor Lb. The fixed time constant capacitor Cb, the winding Ncb and the fixed time constant inductor Lb are connected in series. The generated switching signal is supplied from the first resonance circuit to the base of the switching transistor Q1 so as to control the switching thereof.
The voltage resonance converter 7 further includes a diode D1, a base clamping diode D2 and a film capacitor C1. When the oscillating voltage Vcp1 and the collector current Icp1 change due to the on/off operation of the switching transistor Q1 as illustrated in FIGS. 2(a) and 2(b), respectively, the wave pulse voltage V2 (FIG. 2(c)) generated in the secondary winding Nc2 of the converter transformer CT is applied to the smoothing capacitor Ci by way of the diode D1, and the charge in the smoothing capacitor Ci is repeated by a current I2. See FIGS. 2c and 2d for a representation or voltage V2 and current I2, respectively. When the switching transistor Q1 is not switching, i.e. while the a.c. voltage Vac is low, the base clamping diode D2 functions as a discharge diode.
FIG. 3 illustrates operating wave forms for signals obtained from commercial a.c. power supplies having a frequency of 50 to 60 Hz. The operation illustrated in FIG. 2 is shown in FIG. 3 as the operation within the period t.
The switching transistor Q1 only switches and the voltage resonance converter circuit 7 of the partial rectification smoothing circuit 102 operates as described above, during the period t when the a.c. voltage Vac is relatively high. Furthermore, during this period t, the smoothing capacitor Ci is charged by the a.c. voltage Vac and the voltage V2. Outside this period t, the switching operation of the switching transistor Q1 for the voltage resonance converter 7 is stopped. During this stopped period, the current flows in the direction from the diode D2 to the base of the switching transistor Q1, to the collector of the switching transistor Q1, to the primary winding Nc1 of the converter transformer CT and to the smoothing capacitor Ci, whereupon the capacitor Ci is charged. The charge stored in the capacitor Ci may thereafter be discharged on line 6 as the load current I0.
As a result of the above-described operation of the partial rectification smoothing circuit 102, a direct current (d.c.) rectification smoothed voltage Ei is produced on line 6 from the direct current (d.c.) pulsating voltage received from the full wave rectification circuit 5. However, a ripple voltage component .DELTA.Ei is superposed on this d.c. rectification smoothed voltage Ei on line 6 as shown in FIG. 3(e). As a result, the angle of conduction of the peak current of the a.c. input current Iac, which corresponds to the smoothed d.c. voltage Ei and which is shown in FIG. 3(f) in dotted lines, is expanded. Such current Iac is supplied from the power supply 1 and, as a result, the efficiency of this power supply circuit is improved.
Thus, the partial rectification smoothing circuit 102, which includes the voltage resonance converter 7, operates so as to improve the power efficiency. The switching transistor Q1, which performs the switching operations in the voltage resonance converter so as to control the power efficiency improvement thereof, will hereinafter also be referred to as a resonance switch.
The smoothed d.c. voltage Ei from the partial rectification smoothing circuit 102 is supplied to the last stage of the switching power supply circuit 12. The switching power supply circuit 12 includes a resonance converter circuit (RCC) 9 having switching transistors Q2 and Q3, diodes D3 and D4 which act as base clamping diodes, resistors R2, R3, Rb1 and Rb2, inductors Lb1 and Lb2, and capacitors Cb1 and Cb2.
The windings Lb1 and Lb2 are part of a converter driven transformer (CDT) 10. The winding Lb1 of the converter drive transformer CDT 10, the dumping resistor Rb1 and the time constant capacitor Cb1 are connected in series and coupled to the base of the switching transistor Q2. The winding Lb1 and the time constant capacitor Cb1 constitute a first self resonating circuit. As a result, a first oscillating frequency signal is generated by the first self resonating circuit and supplied to the base of the switching transistor Q2, so as to cause the switching transistor Q2 to operate accordingly.
Similarly, the winding Lb2 of the converter driven transformer CDT 10, the resistor Rb2 and the capacitor Cb2 are connected in series and coupled to the base of the switching transistor Q3. The winding Lb2 and the capacitor Cb2 constitute a second self resonating circuit, which generates a second oscillating frequency signal which is supplied to the base of the switching transistor Q3, so as to cause the switching transistor Q3 to operate accordingly.
The switching transistors Q2 and Q3 perform switching operations in reverse phase to each other.
The switching power supply circuit 12 further includes the CDT 10, a power regulating transformer (PRT) 11, a capacitor C2 and a control circuit 8. In addition to the windings Lb1 and Lb2, the CDT 10 further includes a winding Lb0, which is coupled to the PRT 11. The PRT 11 includes a winding N1, which is coupled to the winding Lb0 of the CDT 10, and windings N2 and Nc.
When the switching transistor Q2 is turned on, and the switching transistor Q3 is turned off, current flows through the collector and emitter of the switching transistor Q2, to the winding Lb0 of the CDT 10, and then to the winding N1 of the PRT 11, so as to charge the capacitor C2. On the other hand, when the switching transistor Q2 is off, and the switching transistor Q3 is on, current flows from the capacitor C2 through the winding N1 of the PRT 11, the winding Lb0 of the CDT 10, and across the collector and emitter of the switching transistor Q3.
FIGS. 2(e), 2(f) and 2(g) respectively illustrate the voltage Vcp3, which is at the collector of the switching transistor Q3, and the currents Icp2 and Icp3, which respectively flow through the switching transistors Q2 and Q3.
Accordingly, by having the switching transistors Q2 and Q3 turn on reciprocally as previously described, an a.c. current is induced at the second winding N2 of the PRT 11. Such current is rectified and smoothed by a rectifying smoothing circuit having diodes D5, D6 and D7, which are coupled to the winding N2, and smoothing capacitors C0 and C3. This rectified and smoothed current may be extracted as a d.c. voltage E0.
The control circuit 8, which is a control error amplifier, includes a zener diode D8 and a transistor Q4. Such control circuit 8 receives the voltage generated in the winding N2 of the PRT 11 by way of resistors R4 and R5 and causes a d.c. control current to flow through the control winding Nc of the PRT 11 due to the zener diode D8 and transistor Q4. As a result, the PRT 11 provides a fixed voltage output.
The individual circuit components without reference designations depicted in FIG. 1 are connected as shown and will not be discussed further, since the connections, values and descriptions thereof are apparent to those skilled in the art and are not necessary for an understanding of the present invention.
In this type of switching power supply apparatus, there are various problems as described below.
First, in the case described above where a partial rectification and smoothing circuit 102 and a switching power supply circuit section 12 are combined, the power efficiency is improved from approximately 0.6 to 0.8 as compared to an apparatus in which a partial rectification smoothing section is not provided. Further, in such apparatus, the high frequency current is reduced. However, as previously described, a ripple voltage .DELTA.Ei is included in the d.c. input voltage Ei for the switching power supply circuit 12. The absolute value of such .DELTA.Ei may be two or three times higher if the partial rectification and smoothing circuit is not provided. The minimum input voltage Ei(min) is Ei(min)=Vm-.DELTA.Ei (see FIG. 3(e)). To maintain the lower limit of the regulation scope, the number of turns on the winding N1 of the PRT 11 should be reduced and the storage capacity of the resonance capacitor C2 should be increased.
In other words, in such apparatus, since the ripple component .DELTA.Ei is large, it may be necessary to provide a circuit for increasing the amount of control sensitivity of the control circuit 8 on the switching power supply circuit section 12 to improve the ripple suppression rate so as to reduce the ripple component of the d.c. output voltage E0 or, alternately, it may be necessary to increase the capacitance of the capacitor C0 or so forth. Further, to maintain the lower limit regulation scope setting, a redesign of the switching power supply circuit 12 may be necessary.
Second, problems may occur from having the switching frequency of the switching power supply circuit 12 and that of the partial rectification smoothing circuit 102 operate independently from each other. For example, the frequency of the signals generated by the first and second self resonant circuits of the switching power supply circuit 12 (i.e., the windings Lb1, Lb2 of the CDT 10 and the fixed time constant capacitors Cb1, Cb2) may be 100 kHz. Whereas, the frequency of the signal generated by the resonance circuit 7 in the partial rectification smoothing circuit 102 (i.e., by the winding Ncb, the fixed time constant inductor Lb, and the fixed time constant capacitor Cb)may be 83.3 kHz. Differences in such switching operation wave forms may be readily apparent by comparing FIGS. 2(a), 2(b) and 2(e), 2(f), 2(g).
As a result, since a resonating circuit is required for both the switching power supply circuit 12 and the partial rectification smoothing circuit 102 (in particular, the voltage resonance converter 7), the number of parts, the amount of substrate surface area and the cost are relatively high. Further, such duplication of resonating circuits increases the difficultly in miniaturizing the circuit. Furthermore, since the switching frequencies are different, the switching power supply circuit and/or the voltage resonance converter may oscillate abnormally due to stray magnetic flux or crosstalk caused, for example, by EMI generated by the reactive elements (inductances, choke coils, transformers and so forth). As a result, the capacitances of the capacitors C1 and C0 may have to be relatively large, or shielding may have to be installed between the elements.
Third, as previously described, the switching frequency of the voltage resonance converter of the partial rectification smoothing circuit is determined by the resonating circuit having the fixed time constant capacitor Cb, and the combined inductance of the winding Ncb and the fixed time constant inductor Lb. As a result, the power efficiency will decrease due to a rise in the a.c. input voltage and a fall in the load current. For example, consider FIGS. 4(a) to 4(f). As shown therein, FIGS. 4(a) to 4(c) and 4(d) to 4(f) respectively illustrate the collector current Icp1 for the switching transistor Q1, the collector current Icp3 for the switching transistor Q3, and the a.c. input current Iac when the switching frequency for voltage resonance converter 7 is 100 kHz and the switching frequency for the switching power supply circuit 12 is 83.3 kHz. However, the wave forms shown in FIGS. 4(a) to 4(c) correspond to a relatively heavy load and an a.c. voltage of 90 V, whereas those shown in FIGS. 4(d) to 4(f) correspond to a relatively light load and an a.c. voltage of 120 V.
As shown in FIG. 4, the switching frequency of the voltage resonance converter 7 of the partial rectification smoothing circuit 102 does not depend on the a.c. input voltage and the current load. That is, in these situations, the switching frequency remains approximately constant. Further, even if the power efficiency is 0.85 when there is a heavy load, the power efficiency drops to 0.6 for light loads due to the narrowing of the conduction angle for the current Iac.
The above-described problems or disadvantages do not only occur when the switching power supply circuit 12 employs a current resonance converter which operates in accordance with a series oscillation frequency control technique and which uses a power regulating transformer, as in circuit 100 of FIG. 1. Such problems may also exist for switching power supply apparatus utilizing other types of converters and/or techniques. For example, such problems may exist for switching power supply apparatus having a current resonance converter which operates in accordance with a switching frequency control technique and uses a power regulating transformer, a voltage resonance converter which operates in accordance with a magnetic flux control technique and uses a power regulating transformer, a flyback converter which operates in accordance with a pulse width modulation (PWM) control technique, a forward converter which operates in accordance with a PWM control technique, a ringing choke converter which operates in accordance with a switching frequency control technique, or the like.
Thus, the prior art has failed to provide a switching power supply apparatus in which ripple components are reduced and in which the aforementioned problems associated with separate resonant circuits in the partial rectification smoothing circuit and the switching power supply circuit of the apparatus are eliminated or minimized.